For many years, power processing and conditioning circuitry and systems used a linear technique to regulate output voltage. Under this technique, the output voltage of a power supply is regulated by deliberate dissipation of some of the energy entering the power supply. The energy is dissipated in a semiconductor device, typically a bipolar junction transistor operating in the linear (active) region. Prior art linear power supplies achieve low operating efficiencies of about 50%. Low operating efficiency is manifested by excessive heating of the semiconductor device. To maintain the temperature of the semiconductor device within prescribed specifications, a large and heavy heat sink is required.
Linear power supplies employ conventional 50/60 Hz power transformers followed by a rectifier, filter and linear regulator. Because of the low frequency of processing, the transformer, as well as filter components, are large and heavy.
Miniaturization of the electronic equipment requires reduction of the size and weight of the power supplies used in that equipment. This is achieved by using switch-mode power supplies with semiconductor devices operating as electronic switches. By principle of operation, there is no deliberate power dissipation in the switching devices. Thus, the switch-mode power supplies are capable of high efficiency. For example, in practical circuits efficiency of 80% is achieved. In switch-mode power supplies the power processing is performed at much higher frequency than in linear power supplies, for example, at 20 or 100 kHz. This results in substantial reduction of the size of the transformer and filter components.
One of the simplest switch-mode DC/DC power converters is the pulse width modulated (PWM) buck converter, shown in FIG. 1A with its operating waveforms shown in FIG. 1B. In FIG. 1B, the first graph shows the on/off states of switch S, the second graph shows the current through switch S, the third graph shows the voltage across switch S, the fourth graph shows the current through the diode D and the fifth shows the voltage across diode D. Switch S is typically implemented using a power metal oxide semiconductor field effect transistor (MOSFET) or bipolar junction transistor (BJT). Inductor L.sub.F and capacitor C.sub.F form an output filter. Resistor R.sub.L represents a load to which power is delivered.
Switch S is periodically closed and open. When S is closed, the input voltage V.sub.IN is applied to the diode D. The diode is reverse biased and does not conduct current. During this time (one-time), energy is delivered to the circuit from V.sub.IN and stored in the filter components L.sub.F and C.sub.F. When switch S is open, the inductor current is diverted from the switch to the diode, and the diode is turned on. During this time (off-time), the filter components release the energy stored during the on-time into the load. The regulation of the output voltage V.sub.O is achieved by varying the ratio of the on-time to the off-time.
Reduction of the size and weight of the switching power supply can be achieved by increasing the switching frequency. However, as the switching frequency increases, so do the switching losses in the switching device. Typically, switching losses increase proportionally to the frequency.
Switching losses can be divided into two categories: turn-off and turn-on losses. The turn-off losses are caused by simultaneous non-zero-voltage and non-zero-current applied to the switching device during turn-off. These losses can be reduced by reducing the turn-off time, which can be achieve by using fast switching transistors and appropriate drive circuits.
Turn-on losses, on the other hand, are caused by dissipation of the energy stored in the parasitic capacitance which exists in parallel with the switching device. This energy is completely dissipated in the switching transistor during turn-on. The turn-on power loss depends only on the energy stored in the parasitic capacitance and switching frequency and is not affected by the switching speed of the device.
Reduction of switching losses also can be achieved by applying certain of the concepts of resonant switching. One concept of resonant switching involves the use of quasi-resonant switches.
Two types of quasi-resonant switches have been introduced: A zero-current quasi-resonant switch as shown in FIG. 2A, and a zero-voltage quasi-resonant switch as shown in FIG. 2B. The two types of quasi-resonant switches are considered in co-pending U.S. patent applications Ser. Nos. 856,775; 877,184 and 877,185, incorporated by reference herein. In the zero-current quasi-resonant switch, the resonant components, L.sub.R and C.sub.R, shape the current through the switch S so that the current becomes zero prior to turn-off of the switch S. This reduces the turn-off losses. In the zero-voltage quasi-resonant switch, the resonant components shape the voltage across the switch S so that the voltage becomes zero prior to turn-on of the switch S.
Any PWM circuit can be converted into a quasi-resonant circuit by replacing the PWM active switch with a quasi-resonant switch. For example, FIG. 3A shows a buck zero-current-switched quasi-resonant converter with its operating waveforms shown in FIG. 2B. The arrangement of the waveforms in FIG. 3B is similar to the arrangement in FIG. 1B. It can be seen that in the zero-current-switched converter, the current through the switch S is quasi-sinusoidal and reduces to zero before switch S is turned off. The turn-on of switch S, however, occurs with the input voltage V.sub.IN applied to the switch. This causes turn-on power dissipation. Maximum switching frequency of zero-current-switched quasi-resonant converters is limited to about 2 MHz.
It can be seen in FIGS. 3C and 3D, that in the zero-voltage-switched quasi-resonant converter the voltage across the active switch is quasi-sinusoidal and reduced to zero prior to turn-on of the switch. This eliminates the turn-on losses. Zero-voltage-switched quasi-resonant converters can operate at frequencies to about 10 to 20 MHz.
Zero-voltage-switched quasi-resonant converters are capable of high operating frequencies. However, they have two major limitations. One problem is excessive voltage stress to the switching transistor. This voltage stress is proportional to the load range which makes it difficult to implement zero-voltage-switched quasi-resonant converters in applications where load varies over a wide range.
Another problem is caused by the parasitic junction capacitance of the rectifying diode D (see FIG. 3C). Theoretical zero-voltage-switched quasi-resonant circuits do not include capacitance in parallel with the rectifying diode. The theoretical voltage across the diode is in the form of a quasi-square wave. In practice, the junction capacitance of the rectifier does not allow for such abrupt changes of the voltage. As a result, parasitic oscillations of the resonant inductance and rectifier's capacitance occur in the circuit. If damped, these oscillations cause power dissipation. If undamped, they adversely effect voltage gain of the converter which makes the converter difficult to control.
There is thus a need for a miniature DC/DC converter capable of operating at high frequency with reduction of voltage stress to the switching transistor, increase of the load range and reduction of the switching frequency range. The present invention is directed toward filling that need.